Dynamic threshold pass-transistor logic for improved delay at lower power supply voltages

A domain partition approach to parallel adaptive simulation of dynamic threshold voltage MOSFET

The four-quadrant analog multipliers (FQAM) are widely used in signal processing applications such as amplitude modulation, frequency doubling, and adaptive filters. In this paper, a FQAM has been realised in dynamic threshold voltage MOSFET (DTMOS) technology. It relies on the square-law characteristics of the MOS transistor. Several simulations have been carried out to assess the operation of the proposed FQAM. The supply voltage is set to ±0.2 V, and 68 µW power consumption is determined. The input signal can be applied to the full scale of the supply voltage. The bandwidth is 22.86 MHz, and the output signal’s total harmonic distortion (THD) is less than 2%. Intermodulation products of the output signal have been calculated. Monte Carlo and temperature simulations have been performed in a way that confirms the robustness of the circuit against the technological spread. In summary, low power consumption, wide bandwidth, and linearity are the advantages of the proposed circuit.

Multiplier is one of the most important arithmetic unit in Microprocessors and DSPs and also a major source of power dissipation. Reducing the power dissipation of multipliers is a key to satisfy the overall power budget of various digital circuits and systems. this paper elaborates the array multiplier through different logic styles. The fundamental units to design a multiplier are adders. The various types of adders used in this paper are complementary MOS (CMOS) logic style, complementary passtransistor (CPL) logic style, double-pass transistor (DPL) logic style and domino logic style. The main objective of our work is to calculate the average power, delay and PDP of 4x4 multipliers. The design of full adder for low power is obtained and the low power units are implemented on the array multiplier and the results are analyzed for better performance. The designs are done using TANNER S-EDIT tool and are simulated using T-SPICE. The multiplier architectures are designed using the three better above said full adders and the results are compared so that we can obtain a better multiplier design.

Reports a novel low-voltage silicon-on-insulator (SOI) CMOS complementary pass-transistor logic (CPL) circuit using asymmetrical dynamic threshold pass-transistor (ADTPT) technique., Using the ADTPT technique to dynamically control the body bias of the pass-transistor via only one auxiliary transistor, the new SOI CMOS complementary pass-transistor logic (CPL) circuit provides superior speed performance at a low supply voltage as compared to the conventional pass-transistor logic circuits without the ADTPT technique as verified by the MEDICI simulation results. The ADTPT technique is especially effective for use in CPL circuits with serially-connected multiple inputs.

In this paper, a low-power high-performance logic style for low-voltage CMOS technologies is presented. The style is based on modifying a high-speed yet low-power logic family called feedthrough logic (FTL) style which has been previously proposed in the literature. The proposed style which is called parallel FTL (PFTL) overcomes the shortcoming of the FTL for low-voltage applications. To assess the efficiency of the proposed logic style, HSPICE simulations are used to compare the performance parameters of PFTL, FTL, and static CMOS logic. The simulation results in 45 nm standard CMOS show a superiority of the proposed structure, especially at low supply voltages. The PFTL has reduced the delay over the standard CMOS logic by 12 to 16 times and over the FTL structure by 3 times, and also PFTL has reduced (delay × power/frequency) ratio over the FTL by 3.5 times and over the standard CMOS by 30 percent at low supply voltages.

Carbon nanotubes (CNT) field-effect transistor (CNTFET) could be a possible alternative to CMOS technology for future VLSI applications. In this work, a comparative study has been carried out on the effects of technology nodes and logic styles on power dissipation, delay, leakages, etc. The technology nodes that are considered here are 90 nm and 32 nm MOSFET technology, and 32 nm CNTFET technology. The logic families considered here are the conventional complementary metal-oxide-semiconductor (CMOS), complementary pass transistor logic (CPL) and transmission gate (TG). The digital circuits considered are NAND, NOR, XOR and MUX gates. HSPICE simulations have been carried out and observed that at 32 nm CNTFET technology, the least power, worst-case delay and least PDP are found as 15.5 nW, 3.11 ps and 0.048 aJ, respectively. It is witnessed that CNTFET-based logics are superior compared to other logic families at different technology nodes.

An ultra-thin natural-source-based cellulose derivate is used in a bilayer gate dielectric for organic complementary inverters. Extraordinarily high small-signal gains up to 1600, a very sharp transition region (as low as 40 mV), a full rail-to-rail swing, a balanced noise margin of 92.5%, and “long-time” stability are shown, revealing that these inverters are highly suitable for complex circuit designs. For more than a decade now the research in organic electronics was driven by unique consumer markets demanding for electronic circuits on flexible substrates over large areas and ideally at low cost. Meanwhile, versatile macroelectronic applications with attractive form factors such as flexible displays,1-4 large-area sensor arrays,5-11 e-paper,12, 13 low-cost radio-frequency identification tags,14-18 and bendable microprocessors19 have been realized using organic thin-film transistors (OTFTs) as the essential element. In the majority of these applications OTFTs are integrated either in the form of arrays or as digital logic. The fundamental building block of digital logic is the inverter, a circuit that inverts an input signal. Much work has been devoted in the last five years to design unipolar inverters using p-type-only OTFTs.18-23 Unfortunately, unipolar logic based on single gate OTFTs always suffers from high power consumption and low noise margin coupled with high noise margin variability and therefore impedes the fabrication of complex circuits having hundreds or more logic gates.19, 22, 23 In this context, it stands beyond doubt that the integration of both n-type and p-type transistors in complementary organic-based circuits12 is very advantageous due to a lower power dissipation, higher noise margin, better operational stability, and easier design of the circuit.24-27 To date, several major obstacles complicated the development of organic complementary circuits, which are as follows: i) the lack of n-channel OTFTs with a performance comparable to the p-channel OTFTs, ii) the challenge of device stability (difficulty in fabricating OTFTs with hysteresis-free characteristics), and iii) the difficulty in fabricating sufficiently thin gate dielectrics allowing for low supply voltages. Although many recently reported complementary organic inverters are impressive with respect to their low operation voltages (≤5 V),27-36 high static gains (>100 V/V),25, 28, 30, 32 and large noise margins (>80%),25, 35, 37, 38 only few of those can combine low-voltage operation with high gain and sufficient immunity against electrical noise.39 Moreover, there have been few investigations of the influence of hysteresis and threshold voltage shifts on the inverter characteristics, which typically gives rise to electrical instabilities and performance limitations of those devices.32, 39 The inverter's noise margin is the most critical parameter, since it directly determines the maximum complexity of circuits.22 Particularly, interesting applications of organic electronic circuits are found in analog electronics that allow for the read-out and processing of signals from printed large-area sensors and form an interface to silicon electronics. Examples are analog-to-digital (A/D) converters, operational amplifiers, and comparators, all including several tens of transistors, thus requiring both high gain and a large noise margin to be functional as well as sufficient processability. It is interesting to note that the majority of organic complementary inverters already mentioned24-34, 36-40 relied on pentacene as the p-type semiconductor (14 incidences), fluorinated copper phthalocyanine (6 incidences), or N,N′-ditridecylperylene-3,4,9,10-tetracarboxylic diimide (5 incidences) as the n-type material. On the contrary, the variation in gate dielectric materials that are used is much higher with an emphasis on hybrid dielectric systems. Hybrid systems allow accounting for both a high dielectric breakdown voltage and an engineered interface to the semiconductor. Moreover, the implementation of biodegradable and biocompatible materials represents a new niche in the organic electronics research, aimed to address the issues of cost and toxicity to humans and environment posed by nowadays electronics, a topic that was recently reviewed elsewhere.41-43 Accordingly, we have demonstrated the usage of a novel hybrid and biodegradable high-k gate dielectric material system based on a cellulose derivative for low-voltage complementary organic circuits with exceptional high noise margins.35 Previous work done in our laboratories demonstrated the fabrication feasibility of complex circuits, where the electrode patterning was performed by a self-aligned photolithography technique.44, 45 This process can easily be transferred to an anodized aluminum (Al2O3) + trimethylsilyl cellulose (TMSC) based bilayer system; the respective results will be published soon. To follow on this preliminary work we have developed an extraordinarily well-performing complementary inverter technology based on a hybrid dielectric composed of a bilayer of alumina (Al2O3) and trimethylsilyl cellulose (TMSC) and pentacene or C60 for the p-channel and the n-channel organic semiconductor, respectively. It outperforms any inverter reported so far with respect to the combination of excellent key parameters, such as record DC gain of above 500 V/V, low operation voltage of only 4 V, and large noise margin up to 92.5%. In order to clarify the origin for this high performance, special attention is given to the investigation of the semiconductor–dielectric interface, the semiconductor morphology, the nature of traps, the influence of mobility levels on the inverter performance and its long-term stability. This article is not meant to report new materials and systems, but rather find explanations and present a thorough analysis of the extraordinary performance of the cellulose derivative (TMSC) dielectric and the TMSC to semiconductor interface in organic field-effect transistors and simple inverter circuits. All OTFTs presented in this work have a bottom gate, top contact device architecture, as illustrated in Figure 1a. The transistor fabrication process is described in detail in the Experimental Section. The semiconducting layer is pentacene for p-type and C60 (fullerene) for n-type conduction. For the gate dielectric, an inorganic (Al2O3)–organic (TMSC) bilayer is chosen with 28 and 30 nm layer thickness, respectively. The dielectric properties of both single-layer TMSC films and bilayer Al2O3 + TMSC films, were investigated by current–voltage (I(V)), capacitance–frequency (C(f)), and capacitance–voltage (C(V)) measurements in appropriate capacitor structures with 72 nm insulator layer thickness of pure TMSC and 28 + 30 nm of bilayer film of Al2O3 + TMSC, respectively. In Figure 1b, the I(V) curves of a representative capacitor formed by a single layer cellulose dielectric is shown, whereas in Figure S1a (in the Supporting Information) the curve with bilayer dielectric is displayed. A comparison of I(V) curves of a single layer and bilayer dielectric is displayed in Figure 1c. At an electric field of 3.0 MV cm−1 (equivalent to ≈22 V) a current density on the order of 10−6 A cm−2 is observed for the single-layer TMSC. A breakdown field of 4.5 ± 0.3 MV cm−1 is extracted from eight appropriate capacitor structures. One representative C(V) measurement is displayed in Figure 1d. Its capacitance of about 29 nF cm−2 corresponds to a dielectric constant of εR = 2.4 ± 0.2 at 1 kHz with an excellent stability at frequencies down to 0.5 mHz (virtually flat dielectric constant and low dielectric losses which are indicative of a high purity and negligible amount of mobile ionic species) as illustrated in Figure 1e.46 In comparison, capacitors based on the bilayer dielectric have three orders of magnitude lower leakage currents (≈10−9 A cm−2) at an electric field of 3.0 MV cm−1 corresponding to ≈17.5 V and an effective capacitance of 53 nF cm−2 at 1 kHz (see Figure S1a,b, Supporting Information). As can be derived from Figure S1c (in the Supporting Information) the permittivity of the bilayer dielectric also shows excellent stability over the tested frequency range (1 mHz to 10 kHz). The measured capacitance value of the dielectric is in good agreement with calculations by assuming a permittivity of 9 and a thickness of 28 nm for the Al2O3 layer as well as a measured permittivity of 2.4 ± 0.2 (Figure 1e) and 30 nm thickness for the TMSC layer (see the Experimental Section). In any OTFT with bottom gate architecture the surface properties of the gate dielectric (surface energy, surface roughness) decisively affect the arrangement and growth of small molecule semiconductors. The surface energy of spin-coated TMSC thin films was determined by contact angle measurements. With a water contact angle of 98.5°, TMSC can be regarded as a hydrophobic, low energy surface material with a total surface energy of γ = 24.2 ± 0.6 mN m−1, separable in a small polar component γP = 1.8 ± 0.5 mN m−1 and a dispersive component γD = 22.4 ± 0.1 mN m−1. The average surface roughness, indicated by the root mean square (rms)–roughness values, was found to be only 0.77 ± 0.1 nm by atomic force microscopy (AFM) measurements. On this smooth and low energy TMSC surface, a small-grain growth mode with average grain sizes between 150 and 300 nm for pentacene (shown in Figure 1a) and 50 nm for C60 was observed by AFM measurements. In Figure 2, typical electrical characteristics of OTFTs with either C60 or pentacene as the semiconducting layer are plotted. Both p- and n-type transistors are based on the inorganic–organic bilayer gate dielectric composed of Al2O3 and TMSC. The output characteristics ID(VDS) are displayed in Figure 2a,b, whereas the transfer characteristics ID(VGS) are shown in Figure 2c,d. Both transistor types display very small hysteresis between forward and reverse voltage sweeps. The output characteristics show clear saturation, but for the pentacene-based OTFT a non-linear behavior of the drain current at very low drain voltages is observable (Figure 2b). Such an S-shape of the output curve has been already observed and reported for many different device configurations and has sometimes been attributed to a lateral field dependent contact resistance.47-55 In Table 1, the critical device parameters Von (onset voltage), Vthr (threshold voltage), S (subthreshold swing), μlin (charge carrier mobility in the linear regime), and μsat (charge carrier mobility in the saturation regime) are compiled and averaged over eight devices for each transistor type. On average, C60-based OTFTs show a charge carrier mobility of 0.7 cm2 V−1 s−1 in the saturation regime, a threshold voltage of 1.3 V, and a small subthreshold swing of 305 mV dec−1. Pentacene-based OTFTs have on average a mean threshold voltage of −4.7 V, a very small subthreshold swing of 225 mV dec−1, and a field-effect mobility of 0.22 cm2 V−1 s−1 in the saturation regime. The gate leakage currents are very low and in the range of 10−10 A. The balanced performance of our p- and n-type OTFTs, as well as the fact that both transistor types strictly behave as normally off transistors at zero applied gate voltage is a good base for integrating them in complementary organic inverters. A schematic image of such a complementary inverter circuit is displayed in Figure 3a together with the voltage transfer characteristics (VTCs) of a typical device and the static gain measured for operation voltages of VDD = 4, 5, 6, and 7 V. The inverter exhibits extraordinarily high small-signal gains; for all operation voltages, gains above 500 V/V are measured with a maximum value of about 1000 V/V for VDD = 7 V. The best performing device even has a maximum gain value of 1600 V/V. Moreover, a very sharp transition regime (ΔVin ≤ 40 mV) and a full rail-to-rail swing are observed. The extracted inverter parameters are compiled in Table 2. The switching threshold voltages VM are found to be very close to the ideal value of VDD/2 at all operation voltages with a nearly perfect fit (e.g., VM = 2.04 V at VDD = 4 V). In Figure 3b, the mismatches of the threshold voltages to the ideal values (ΔVM) are plotted as a function of VDD revealing a linear increase of ΔVM with the supply voltage. This means that the asymmetry of the VTCs increase with increasing the operation voltages. Additionally, the noise margins (NM) are plotted in Figure 3b as a function of VDD, illustrating the linear decrease of the NM with increasing operation voltage. Such a behavior reveals the direct correlation between ΔVM and NM and is an indication of the deviation of the VTCs from perfect symmetry. The noise margin is an important figure of merit of an inverter stage and measures the immunity against inevitable transistor parameter variations and electrical noise.56 There are several ways to determine the noise margin of an inverter. One convenient way is called the “maximum equal criterion” (MEC), where the NM is found by the maximum square that fits between inverter transition curve and the mirrored inverter curve as shown in Figure 3c.22, 35 The NM values for all the supplied voltages are listed in Table 2. The noise margin is mainly limited by the steepness of the transition ΔVin and the position of the switching threshold voltage VM (VM = Vin = Vout) as shown in Figure 3c. High and balanced noise margins are found when the ideal condition VM = VDD/2 is met, thus when VM = Vin =VDD/2 = Vout. The inverter in Figure 3c has a NM of 1.79, which is 90% of the maximum theoretical value for a supply voltage of 4 V, determined by the “maximum equal criterion.” Additionally, the “negative slope criteria” reveals a balanced NM of 1.85 V (92.5%) for a supply voltage of 4 V. The OTFTs with pentacene and C60 semiconductor layers (as used in the complementary inverters) were also analyzed with regard to their long-term stability. The dependence of their onset voltage and charge carrier mobility on the time period after preparation is plotted in Figure 4 based on measurements of at least five devices for each transistor type. The error bars display the standard deviations. p- and n-type devices exhibit an onset voltage shift of +1.3 and +0.95 V, respectively, over a period of 45 days. The main part of the shift occurred either after the first or after the second measurement and then the onset voltage shift tended to saturate. It seems that both pentacene- and fullerene-based devices stabilized their electrical properties after few days. Between day 8 and day 45 no significant change in the transfer characteristics was observed. Concerning the subthreshold swing, no significant change was observed over 45 days (S = 215 mV dec−1 (0 days), 210 mV dec−1 (8 days), and 225 mV dec−1 (45 days)) in the pentacene OTFTs, indicating that the interface trap density is stable. On the contrary, after 8 days an increase of the swing was observed for the C60 devices from 305 mV dec−1 (0 days), to 410 mV dec−1 (8 days), to 430 mV dec−1 (45 days). The mean value of the charge carrier mobility of the pentacene OTFTs showed no significant variation over 45 days: 0.22 cm2 V−1 s−1(fresh sample), 0.19 cm2 V−1 s−1 (after 8 days), 0.21 cm2 V−1 s−1 (after 45 days). In contrast, a strong reduction of about one order of magnitude (from 0.7 to 0.06 cm2 V−1 s−1 within the first 8 days and to 0.05 cm2 V−1 s−1 after 45 days) was observed for the fullerene transistors. In order to test the reliability of our OTFT devices we have also performed negative and positive bias stress tests. The bias stress was done 1 day after fabrication, on devices which had been previously measured only once. A reversible negative onset voltage shift of about 400 mV was observed under negative bias of −7.5 V over 21 hours stress (see Figure 5a,b) The main part of the negative voltage shift occurred during the first 2 hours (300 mV) whereas shift saturation was reached after 6 hours. The positive bias stress resulted in a significantly smaller positive onset voltage shift (ca. +100 mV over 4 hours stressing the device). Devices that were stored for seven months in the glovebox showed no bias stress effect at all (i.e., shift of the onset voltage) when their transistor characteristics were measured at regular intervals during the storage period. In order to investigate the “long-time” behavior of complementary inverters, electrical measurements were done after 1, 4, 8, and 45 days of the respective device preparation. Inverters were measured 1 day after device stabilization (see Figure 4a,b). The “long-time” behavior of complementary inverters is shown in Figure 4c,d where the VTC and gain curves are plotted as a function of time. For the lowest VDD of only 4 V, a small shift of the switching threshold voltage was measured: starting from 2.04 V after 1 day to an identical value of 2.17 V after 8 and 45 days, respectively, with a related exceptional high noise margin of 1.79 V (90% of the theoretical maximum) after 1 day and 1.74 V (87%) after 8 and 45 days. It is interesting to note that, after 8 days of storage time, the hysteresis in inverter characteristics was virtually negligible (<100 mV, see Figure 4d) and moreover, repeated measurements did not change the inverter transition, although the gain was somewhat reduced (from 503 (0 days) to 320 (8 days) and 195 (45 days)). The operational stability of the VTC was shown over a period of at least 45 days (Figure 4c for VDD = 4 V). Furthermore, the remarkable long-term stability of these TMSC-based complementary inverters was demonstrated by 16 consecutively recorded inverter curves that are virtually superimposed over one another (Figure 5c). The current–voltage characteristics show that the Al2O3 + TMSC bilayer capacitors have substantially lower leakage currents than the single TMSC layer devices. Due to the roughness of the aluminum bottom electrode with spikes up to 10 nm, the anodization process is a convenient method to reduce the leakage currents for at least three orders of magnitude at an electric field of 3 MV cm−1 as displayed in Figure 1d and make the dielectric layer more robust.57 Another advantage of the anodization step is the substitution of the hydroxide of the native oxide layer. Finally, the subsequent short oxygen plasma treatment (see the Experimental Section) is sufficiently removing all residuals left after the anodization process. This procedure allows for the generation of a pure and trap-reduced dielectric interface to the organic semiconductors, by measurements at low revealing a virtually flat permittivity in the mHz to kHz frequency range and low dielectric losses which both are indicative of a high purity and negligible amount of mobile ionic The generation of this excellent interface is in virtually hysteresis-free characteristics of all the One important that the performance of OTFTs, and in the charge carrier is the of the As shown by several the of the surface energy of the to that of the is one important the growth For a low surface energy material with a smooth surface TMSC, a growth mode is and with a rather high density and small grain sizes are formed (see Figure The high density of grain a significant decrease of the charge carrier since grain as charge On the a better between pentacene on low surface energy materials TMSC with and better reduce the of trap thus the mobility of charge in the In there is an between grain related and interface with regard to their influence on charge carrier The low subthreshold swing observed in all our pentacene-based OTFTs is indicative of a low interface trap density which the influence of the large grain trap density on the charge carrier Accordingly, a mobility 0.2 cm2 V−1 s−1 is observed in the p-type devices. In C60 films, the influence of on the charge carrier mobility is by far not as significant as in pentacene films, which is attributed to the of the fullerene In the pentacene OTFTs (see Figure a non-linear current increase in the linear regime of the ID(VDS) output characteristics is at higher gate have been in to such a which can be observed in bottom gate The explanations i) the of the charge carrier mobility for semiconductor ii) currents that the of the mobile by an effective under the drain and iii) large at the electrode from an energy between the work of contact material and semiconductor films, a gate or large contact are found in our of these it has been shown that a mobility in combination with a small contact in a non-linear drain voltage dependence of the current due to The from a of from trap and a dependence of on the drain A respective of the pentacene output curve in Figure the dependence in the regime 2 V, thus indicating that a mobility has to be The of any hysteresis in the I(V) curves and the low and value of the subthreshold swing a low density of interface in the pentacene are not for the observed and the of the as by the of the pentacene layer gives rise to which to a field dependence of the charge carrier This is all the more given that the pentacene grain sizes are 300 nm at a large density of grain As by Figure 4 the characteristics of both pentacene- and C60-based OTFTs show a positive onset voltage shifts storage which after a of measurements. The virtually subthreshold swing in pentacene as well as the negligible hysteresis as the origin of the On the contrary, it seems that trap are for the positive onset voltage shift in are either over time or during the first measurements by the applied positive gate be easily by the gate bias and therefore are not to the swing, but the small onset voltage in pentacene-based It has been reported by many that the of the of trap in pentacene trap also be due to such as at the a at the or a at the fabrication and transfer from the to the measurement stage the semiconductor layer and rise to such For oxygen can even the devices are to but transferred directly and measured in a glovebox with reduced oxygen and water have also observed that the of oxygen with pentacene is in the that these are under device operation with the of oxygen in the grain observed that the storage of devices in and their subsequent measurement in did not a change in their I(V) also observed that the storage time to the first measurement has influence on the device (i.e., on the onset voltage in this of and the first measurement the of to the of oxygen or with the which in a second are by Moreover, also are are not as in the as the (see and This an for the positive shift of the onset voltage between first and second measurement (Figure and also for the saturation of the onset voltage shift in In order to clarify which of are present in our we performed measurements on pentacene films (in and after storage in for (see Figure Supporting Information). storage under a small increase of the at energy is observed that can be attributed to the to and The analysis reveals an increase of the from 0.2 to At the position no change is that trap in our devices for bias stress and the positive onset voltage can mainly be attributed to It be that the of any mobility in pentacene OTFTs against an of pentacene both are to oxygen For the C60 transistors, one a strong decrease of the charge carrier mobility during 8 days of storage which is by all over the device grain This is a effect of The decrease in charge carrier mobility and the at the interface that the subthreshold swing is in a positive onset voltage shift of the C60 transistors. With respect to the complementary inverters, noise margin values beyond from the maximum theoretical value for all supply voltages are and the high operational of the OTFT devices. At a very low supply voltage of 4 V, our bilayer organic complementary inverters have a static gain of more than 500 V/V with a noise margin of 92.5%, which is already reported devices with architecture (see the of inverter performance given in 38 and also gain values above 1000 V/V values up to 1600 were measured for our devices. This is all the most as the charge carrier mobility values of the transistors are not thus the that the of the mobility values between p- and n-channel with the virtually hysteresis are the factors for high gain and high noise margin Accordingly, a mobility 1 be the the is not high ≤ it can be by and p-channel transistors with different In our the mobility is The static of a complementary inverter is determined by technology parameters, the parameters of p- and n-type transistors. A good of these parameters a good saturation of both output can to a significant increase of the gain A is shown in the Supporting Furthermore, high gain values are also determined by the interface An insulator with an low trap density (as this is the for is in having I(V) curves with virtually zero Accordingly, it is to measurements with a very high voltage small voltage thus one to the high gain value of the complementary inverter it by hysteresis The noise margin with increasing VDD due to their strong this is also for the of the VTCs by see Figure A for this behavior can be found in the bias stress that was observed for both types of transistors. The negative shift of Von under negative gate and the positive

As feature sizes of integrated circuits continue to shrink and demand for low power operation continues to increase, power supply voltages may eventually be reduced to the level where noise becomes significant compared with signals. This would cause digital logic circuits to exhibit non-deterministic behavior. In this paper, we study the stochastic behavior of simple digital circuits operating at low power supply voltages. We develop a method for modelling the stochastic behavior of circuits and apply the method to a CMOS inverter. We use numerical simulations of stochastic differential equations to obtain time- domain transient analysis of the CMOS inverter and magnitude statistics from multiple sample paths. Further, we derive the output distribution at steady state from first principles. We characterize the switching error and the information transmission based on the output steady state distribution. This study provides a detailed understanding of how noise affects the operation of the CMOS inverter at low power supply voltages. The method we develop here can also be extended to other circuits where stochastic sample paths and sample statistics are necessary to characterize the circuits. Traditional frequency domain noise analysis is not adequate to provide this information.

Improvement of diagnosis methodologies is a key factor for fast failure analysis and yield improvement. As bridging defects are a common defect type in CMOS circuits, diagnosing this class of defect becomes relevant for present and future technologies. Bridging defects cause two additional current components, the bridge and the downstream current. This work presents the effect of the downstream current on current signatures and its impact on the diagnosis of such defects. The authors demonstrate that the impact of downstream current is minimized at low power supply (V <sub xmlns:mml="http://www.w3.org/1998/Math/MathML" xmlns:xlink="http://www.w3.org/1999/xlink">dd</sub> ) values. Therefore, current measurements at low power supply voltages are proposed to enhance bridge diagnosis. Experimental evidence of this behaviour is presented for real devices. Furthermore, current signatures measured at V <sub xmlns:mml="http://www.w3.org/1998/Math/MathML" xmlns:xlink="http://www.w3.org/1999/xlink">VLV</sub> are used for the diagnosis of fifteen failing 0.18mum technology devices, which are demonstrated to contain a bridging defect.

There is a growing need to analyze and optimize the stand-by component of power in digital circuits designed for portable and battery-powered applications. Since these circuits remain in stand-by (or sleep) mode significantly longer than in active mode, their stand-by current, and not their active switching current, determines their battery life. Hence, stringent specifications are being placed on the stand-by (or leakage) current drawn by such devices. As the power supply voltage is reduced, the threshold voltage of transistors is scaled down to maintain a constant switching speed. Since reducing the threshold voltage increases the leakage of a device exponentially, leakage current has become a dominant factor in the design of VLSI circuits. In this paper, we describe a method that uses simultaneous dynamic voltage scaling (DVS) and adaptive body biasing (ABB) to reduce the total power consumption of a processor under dynamic computational workloads. Analytical models of the leakage current, dynamic power, and frequency as a function of supply voltage and body bias are derived and verified with SPICE simulation. Given these models, we show how to derive an analytical expression for the optimal trade-off between supply voltage and body bias, given a required clock frequency and duration of operation. The proposed method is then applied to a processor and is compared with DVS alone for workloads obtained using real-time monitoring of processor utilization for four typical applications.

Scaling supply voltage is an efficient technique to achieve low energy delay product (EDP). This paper investigates static CMOS, DCVSL, CPL and TG logic full adders in sub-threshold and super-threshold region in terms of low EDP. All circuits are simulated with HSPICE at a PTM 65nm CMOS technology by varying supply voltages from O.2V to 1.1 V with O.1V steps. The simulation results demonstrate that lowering supply voltage is advantageous, especially in medium-voltage region (700mv–800mv) which yields the best EDP. In addition, it is shown that the optimum supply voltage of the full adders varies slightly with logic style.

Parallel Adders or Ripple Carry Adders (RCA) are the building blocks of the digital processing units. However these RCAs have disadvantage of large propagation delay because of the propagation of carry from the first adder to the last adder. To overcome this drawback, Carry Look Ahead Adders (CLAA) are used. But again these CLAAs consumes more power because of use of large number of gates for generating carry. In this paper, we present a design of CLAA based on hybrid CMOS logic style which consumes less power as compared to the designs of CLAA based on complementary CMOS, Pass Transistor Logic (PTL) and Transmission Gate (TG). The proposed design of CLAA has less propagation delay as compared to its existing designs. Also the number of transistors required for the proposed design of CLAA are less as compared to the existing designs based on other logic style.

At present the processing power of the digital electronic chip is enormous and that has been possible because of the continuous improvement of the design methodology and fabrication technology. So, the data processing capability of the chip is increased significantly. Data processing in the electronic chip means the arithmetic operation on that data. For that reason, ALU is present in any processor. Full adder is one of the critical components of arithmetic unit. Improvement of the full adder is necessary for improving the computational performance of a chip. In order to design an efficient full adder, designer should choose an appropriate logic style. In this research, two new model of full-adder circuits are designed and analyzed using Pass Transistor logic in order to reduce power consumption and increase operational speed. The first proposed adder consists of 8 transistors and the second one consists of 10 transistors. LTSPICE is employed for simulating the proposed circuits using16nm low power high-k strained silicon transistor model. The overall performance of the proposed adder circuits and comparative results demonstrate the superiority of the proposed model.

The performance of semiconductor transistors is significantly influenced by charge trapping at oxide and interface defects. The impact of charge-trapping events of defects on the characteristics of the transistor is strongly dependent on factors such as the geometry and the operating point at which the transistor is used. Understanding the complex relationships between the influence of defects and the robustness of devices is essential to optimize circuit performance and becomes particularly important in analog designs. In this work, we investigate the influence of gate oxide defects on the reliability of nanoscale MOS transistors under varying body bias conditions. Using measure-stress-measure techniques, we observe notable effects on both time-zero and time-dependent variability with the application of body bias. Furthermore, the amplitudes of the step heights are investigated as they provide an important measure in scaled technologies to estimate the impact of traps on the device behavior. The results indicate that a body bias can be strategically employed to enhance device reliability by fine-tuning the body bias conditions. • Role of Defects in Device Variability : a. The study focuses on charge trapping at oxide and interface defects, which influence transistor characteristics, such as variability and reliability; b. Analysis highlights how defects manifest as discrete current steps (random telegraph noise, RTN) in nanoscale devices. • Body Bias as a Control Parameter : a. The application of body bias significantly aKects time-zero variability and time-dependent behavior of defects; b. Forward body bias (FBB) reduces variability by modulating the channel depletion region, while reverse body bias (RBB) increases it, providing design flexibility for specific applications. • Experimental Methodology a. An extended measure-stress-measure (eMSM) technique was employed for precise defect analysis; b. Measurements were conducted on nMOS and pMOS devices, revealing diKerent impacts of BB on each type. • Statistical Analysis of Defects and Step-Height Distributions a. Step heights of threshold voltage shifts were statistically analyzed, showing distinct distributions for nMOS (bimodal) and pMOS (unimodal) devices; b. A strong correlation between BB and defect impact was observed, with forward bias reducing and reverse bias enhancing variability. • Benefits for Analog Design a. The BB and FBB can be strategically used to optimize device reliability and circuit performance by tuning threshold voltage variability and reducing noise;

The integrated systems today require flexibility, performance and reconfigurability. The trends in this domain lead to integrate on a single chip different processing cores, communication units and reconfigurable logic. Therefore the Systems on Chip (SoC) can embed programmable logic. In order to challenge the reconfigurability paradigm for special issues such as communication, synchronization or security, the asynchronous logic is a very promising approach. Nevertheless, the standard programmable logic blocks are not well-suited to map asynchronous circuits. The goal of this study is to define a more adequate programmable structure to implement asynchronous designs on SoCs embedding a reconfigurable part. This work is part of a larger project which includes the design of an embedded Programmable Logic Device (e-PLD) dedicated to the implementation of clockless circuits. The more robust and reliable asynchronous circuits are quasi-delay insensitive. These circuits are mainly constructed with Muller gates. The paper presents a new Look-Up Table (LUT) architecture well-adapted to the Muller gate implementation. This new LUT allows the combination of a single memory-point with combinational logic. This programmable memory is realized thanks to an optional feedback structure. This architecture has been evaluated in CMOS, Pass-Transistor Logic and 3-state logic which is a non-conventional way to design LUTs. The simulations report detailed comparisons between the different logic styles and demonstrate for equivalent power consumption a higher speed for 3-state logic.